1. Fie1d of the Invention
This invention relates generally to voltage-or current-regulating systems, and more particularly to regulating systems for spacecraft solar panels or like power sources in a series array.
2. Description of the Related Art
(a) Dissipative Shunt Limiters--Conventional regulating systems for spacecraft solar panels use full-shunt or tap-shunt dissipative voltage limiters as illustrated in FIGS. 4 and 5 respectively.
In these sytems an "error amplifier" compares a calibrated fraction of the output voltage to a reference voltage, and if the output voltage is excessive opens a shunt path that draws current from the solar-panel array (FIG. 4) or from part of it (FIG. 5). As progressively higher current is drawn from the shunted solar panels, the inherent current-voltage characteristic of those panels forces their output voltage downward. If the overvoltage is high enough, the voltage on some of the shunted panels collapses entirely.
In spacecraft these systems are disadvantageous because the shunt current heats the shunt-path transistor switch, and any other elements in the shunt path. This localized heating in turn produces a thermal-balance problem whose magnitude is highly variable with the condition of the solar panels, the operating mode of the load, the sunlight available for collection by the solar panels, and the temperature of the spacecraft--historically as well as instantaneously. In spacecraft that are subject to attitude variations, this parameter too affects the magnitude of the thermal-balance problem. All of these parameters typically vary greatly during and over the life of the craft, making a simple compensating system very difficult to design for all conditions.
The amount of power dissipated or "dumped" in the shunt path typically may be as high as seventy watts in a three-hundred-watt supply system, or roughly twenty-three percent. This is a rather large fraction of the overall dissipation in a spacecraft to be dissipated in a highly localized fashion.
The power dissipation in the full-shunt configuration (FIG. 4) is even larger. Unfortunately, however, when used in the tap-shunt configuration (FIG. 5) dissipative shunt limiters have yet another disadvantage: the selection of the tap-point position along the series array is a very "fussy" design decision. The tap-point location is a critical parameter because the precise amount of the dissipation depends upon that location.
Nevertheless dissipative tap shunt limiters have been considered the best compromise because at least they do not interfere with power transfer from the source to the load when the solar panels are at the end of their useful life. At that time the limiters in effect remove themselves from the circuit, allowing the from the failing panels. load to wring every last small amont of remaining power
(b) Boost Switching Regulators--Boost switching regulators of several types are common in many basic power-handling situations. As in the dissipative shunt limiters discussed above, a boost regulator uses an "error amplifier" to compare some calibrated fraction of the output voltage to a reference voltage; the error amplifier adjusts the output voltage to a generally constant value. Unlike the dissipative limiters, however, a boost regulator does not dump overvoltage through a dissipative auxiliary shunt path.
Rather, a boost regulator operates by continuously controlling the voltage boost ratio through a switched-inductor circuit. The voltage boost ratio of such a circuit is produced inductively as follows.
In the well-known regulator shown in FIG. 6, the transistor switch Q1 cycles on and off. During a first part of the cycle it is turned on, drawing current through the inductor L1 and establishing a magnetic field in the inductor. During a second part of the cycle the switch Q1 is turned off, effectively placing the inductor in series with the power source. The collapse of the magnetic field within the inductor produces a voltage across the inductor, and the resulting current in the inductor flows through the diode D1 into the load.
During this second part of the cycle the voltage applied to the load is equal to the sum of the supply voltage and the inductor voltage. Hence, the circuit can produce a substantial voltage boost ratio. An output capacitor C1 stores the higher voltage, and the diode D1 prevents reverse current from the capacitor back into the inductor or the switch during the first part of each subsequent cycle (when the input end of the diode is essentially grounded). The capacitor also acts as a filter to reduce ripple voltage at the load resulting from the cycling of the switch Q1 and the inductor L1.
The voltage boost ratio of this circuit is controlled continuously by variation of the waveform supplied to the transistor switch Q1. The control signal applied to the base of the switch is a rectangular pulse of variable frequency or duty cycle, or both.
In one common type of system the frequency is fixed while the duty cycle varies; in other words, the range of possible output signals from the error amplifier is converted to a corresponding range of pulse widths in the "pulse-width modulator" block PWM. For example, the modulator PWM can be arranged to lengthen the duration of output-current pulses from the inductor L1 to the load when the load voltage is inadequate, and conversely. In this way the load voltage is servocontrolled to the standard desired value.
Another way of controlling the pulse-width fraction (or duty cycle) is to let the error signal directly control the cycling of the switch Q1 on and off, rather than only controlling the pulse width at a fixed frequency. In some systems of this second type the result is to maintain the absolute pulse width relatively constant while allowing the overall period to vary. In other systems of this same general type, both the pulse width and period vary. In designing either of these subtypes care is required because the system is subject to frequency changes.
Ripple filtering requirements at the regulator output can be substantially reduced by using a two-phase regulator such as illustrated in FIG. 7. This circuit is essentially two of the basic boost-regulator circuits of FIG. 6 placed in parallel and operated in opposed phase. In principle the capacitor C2 in this circuit need handle only half the ripple current.
In both boost-regulator circuits discussed so far, it is possible to conceptualize the function of each inductor as alternating between performing the functions of a transformer primary and a transformer secondary: first the inductor receives a "chopped" voltage or a. c. square wave from the source, then its position in the circuit is in effect switched to deliver a corresponding square wave to the load. These two functions are in effect separated in another boost-regulator variant, the "flyback regulator," appearing in FIG. 8.
Here, as before, an inductor L2 receives the chopped input by operation of the cycling of a transistor switch Q2, but the inductor L2 is coupled to another inductor L3. Loosely speaking, the inductor L2 functions more nearly as the primary of an actual transformer--whose secondary is the other, coupled inductor L3. The latter delivers an output rectangular wave to the load and to the storage and ripple-filtering capactior C3, through a diode D2 which operates as a half-wave rectifier. In this circuit the output voltage from the output inductor L3 is not added to the voltage from the supply; rather the output current from the output inductor L3 is forced into the load in parallel with the current from the supply.
Yet another boost-regulator variant, the "current-fed inverter," appears in FIG. 9. This circuit is in effect a transformer-coupled version of the boost regulator of FIG. 6: it provides separate primary windings L4a and L4b--which are energized in opposed phase--and corresponding separate secondary windings L5a and L5b whose rectified outputs are connected in parallel to provide a two-phase or full-wave output. (Here the interconnected inductors L4a, L4b, coupled to the interconnected inductors L5a and L5b, form a transformer literally.) The primary L4a, L4b is fed through a series inductor L6 which functions generally in the same fashion as the inductor L1 of FIG. 6.
To some people skilled in the art of power electronics, a flyback regulator or a current-fed inverter may not be a species of boost switching regulator. For definiteness of this document we therefore define the phrase "boost switching regulator" to include flyback regulators, current-fed inverters, and in fact any device than can function as a feedbackcontrolled "dc transformer." A feedback-controlled dc transformer is a circuit that (1) accepts dc input power and produces dc output power, and (2) has a variable ratio of output voltage to input voltage, and (3) has a ratio of output current to input current that varies approximately in inverse proportion to the voltage ratio, neglecting internal losses, and (4) servocontrols either of those ratios to hold a parameter of the output power constant.
In all of the boost-regulator circuits illustrated in FIGS. 6 through 9, the power-input terminals are connected across substantially the entire power supply. The power-output terminals are connected across the load, and the voltage-sensing terminals receive a voltage which is a measure (typically a calibrated fraction) of the voltage across the load.
As conventionally used these boost-regulator circuits all share a common disadvantage, namely that all of the power supplied from the power source to the load flows through the boost-regulator components: the inductor or inductors, transistor switch or switches, diode or diodes, and (particularly in the half-wave variants) the capacitor. Since none of these components are ideal, they all have some resistive character and consequently dissipate some energy in the performance of their functions.
Consequently, each boost-regulator circuit has some overall inefficiency, generally between five and ten percent, in its power-handling behavior. In other words, typically five to ten percent of the power flowing from the source to the load is lost in the regulator.
Unfortunately, this dissipation continues unabated during the entire life of the craft. Boost switching regulators as conventionally used therefore waste a very significant fraction of the power from the solar panels of a spacecraft at the end of the spacecraft life, the very time when power waste cannot be tolerated. In effect, boost regulators as so used shorten the overall life of the entire craft--by some fraction related in a complex way to the inefficiency factor of five to ten percent.
(c) Efficiency Comparisons--During tne early part of the useful life of a solar panel, such losses from a boost switching regulator are generally acceptable. The localized heating produced is only about twenty to forty-five percent of the earlier-mentioned power fraction (i.e., twenty-three percent of the overall system power) typically dissipated by dissipative shunt limiters.
However, there is a countervailing consideration. The dissipation in a boost switching regulator arises as a fraction of the total power to the load and therefore continues during the entire life of the craft, whereas the dissipation by dissipative shunt limiters arises as a higly variable fraction of the excess power from the source and therefore vanishes at the end of the life of the spacecraft.
In the absence of the present invention, troublesome thermal-control considerations consequently must be traded off on an all-or-nothing basis against crucial power-availability considerations, in the design of systems for spacecraft power regulation. Heretofore it has been considered necessary to operate spacecraft under either (1) the thermal-control handicaps associated with dissipative shunt limiters or (2) the life-shortening handicaps associated with boost regulators.